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SA5212A Transimpedance amplifier (140MHz)
Product specification Replaces datasheet NE/SA/SE5212A of 1995 Apr 26 IC19 Data Handbook 1998 Oct 07
Philips Semiconductors
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
DESCRIPTION
The SA5212A is a 14k transimpedance, wideband, low noise differential output amplifier, particularly suitable for signal recovery in fiber optic receivers and in any other applications where very low signal levels obtained from high-impedance sources need to be amplified.
PIN CONFIGURATION
N, FE, D Packages
1 2 3 4 8 7 6 5
IIN VCC
GND2 OUT (-) GND2 OUT (+)
FEATURES
GND1 GND1
* Extremely low noise: 2.5pA/Hz * Single 5V supply * Large bandwidth: 140MHz * Differential outputs * Low input/output impedances * 14k differential transresistance * ESD hardened
APPLICATIONS
SD00336
Figure 1. Pin Configuration
* Fiber-optic receivers, analog and digital * Current-to-voltage converters
ORDERING INFORMATION
DESCRIPTION 8-Pin Plastic Small Outline (SO) Package 8-Pin Plastic Dual In-Line Package (DIP) 8-Pin Ceramic Dual In-Line Package (DIP)
* Wideband gain block * Medical and scientific instrumentation * Sensor preamplifiers * Single-ended to differential conversion * Low noise RF amplifiers * RF signal processing
TEMPERATURE RANGE -40C to +85C -40C to +85C -40C to +85C
ORDER CODE SA5212AD SA5212AN SA5212AFE
DWG # SOT96-1 SOT97-1 0580A
ABSOLUTE MAXIMUM RATINGS
SYMBOL VCC Power Supply Power dissipation, TA=25C (still air)1 PD MAX 8-Pin Plastic DIP 8-Pin Plastic SO 8-Pin Cerdip IIN MAX TA TJ TSTG Maximum input current2 Operating ambient temperature range Operating junction Storage temperature range 1100 750 750 5 -40 to 85 -55 to 150 -65 to 150 mW mW mw mA C C C PARAMETER SA5212A 6 UNIT V
NOTES: 1. Maximum dissipation is determined by the operating ambient temperature and the thermal resistance: 8-Pin Plastic DIP: 110C/W 8-Pin Plastic SO: 160C/W 8-Pin Cerdip: 165C/W 2. The use of a pull-up resistor to VCC, for the PIN diode, is recommended
1998 Oct 07
2
853-1266 20142
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
RECOMMENDED OPERATING CONDITIONS
SYMBOL VCC TA TJ Supply voltage range Ambient temperature ranges Junction temperature ranges PARAMETER RATING 4.5 to 5.5 -40 to +85 -40 to +105 UNIT V C C
DC ELECTRICAL CHARACTERISTICS
Minimum and Maximum limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V and TA=25C1. SYMBOL VIN VO VOS ICC IOMAX IIN IN MAX PARAMETER Input bias voltage Output bias voltage Output offset voltage Supply current Output sink/source current Maximum input current (2% linearity) Maximum input current overload threshold Test Circuit 6, Procedure 2 Test Circuit 6, Procedure 4 20 3 40 60 26 4 80 120 TEST CONDITIONS Min 0.55 2.5 Typ 0.8 3.3 Max 1.05 3.8 120 33 UNIT V V mV mA mA A A
NOTES: 1. As in all high frequency circuits, a supply bypass capacitor should be located as close to the part as possible.
1998 Oct 07
3
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
AC ELECTRICAL CHARACTERISTICS
Minimum and Maximum limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V and TA=25C5. SYMBOL RT RO RT RO PARAMETER Transresistance (differential output) Output resistance (differential output) Transresistance (single-ended output) Output resistance (single-ended output) TEST CONDITIONS DC tested, RL = Test Circuit 6, Procedure 1 DC tested DC tested, RL = DC tested Test Circuit 1 D package, f3dB Bandwidth (-3dB) TA = 25C N, FE packages, TA = 25C RIN CIN R/V R/T IN Input resistance Input capacitance Transresistance power supply sensitivity Transresistance ambient temperature sensitivity RMS noise current spectral density (referred to input) Integrated RMS noise current over the bandg width (referred to input) CS = 01 IT CS = 1pF VCC = 5 0.5V D package TA = TA MAX-TA MIN Test Circuit 2 f = 10MHz TA = 25C TA = 25C Test Circuit 2 f = 50MHz f = 100MHz f = 200MHz f = 50MHz f = 100MHz f = 200MHz Any package DC tested VCC = 0.1V Equivalent AC Test Circuit 3 Any package f = 0.1MHz1 Test Circuit 4 RL = Test Circuit 6, Procedure 3 Test Circuit 5 Test Circuit 5 1.7 100 70 120 110 10 9.6 0.05 2.5 20 27 40 22 32 52 nA 150 18 pF %/V %/C pA/Hz 100 140 MHz Min 9.0 14 4.5 7 Typ 14 30 7 15 Max 19 46 9.5 23 UNIT k k
PSRR
Power supply rejection ratio2
20
33
dB
PSRR
Power supply rejection ratio2 (ECL configuration) Maximum differential output voltage swing Maximum input amplitude for output duty cycle of 50 5%3 Rise time for 50mV output signal4
23
dB
VO MAX VIN MAX tR
3.2 325 2.0
VP-P mVP-P ns
NOTES: 1. Package parasitic capacitance amounts to about 0.2pF. 2. PSRR is output referenced and is circuit board layout dependent at higher frequencies. For best performance use RF filter in VCC line. 3. Guaranteed by linearity and over load tests. 4. tR defined as 20-80% rise time. It is guaranteed by -3dB bandwidth test. 5. As in all high frequency circuits, a supply bypass capacitor should be located as close to the part as possible.
1998 Oct 07
4
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TEST CIRCUITS
SINGLE-ENDED Rt + V OUT V IN 2 @ S21 @ R DIFFERENTIAL Rt + V OUT V IN 4 @ S21 @ R
1 ) S22 RO + ZO * 33 1 * S22
1 ) S22 R O + 2Z O * 66 1 * S22 SPECTRUM ANALYZER
NETWORK ANALYZER VCC AV = 60DB 1F
OUT S-PARAMETER TEST SET PORT 1 PORT 2 NC IN DUT
33
33 OUT
1F
VCC GND1 0.1F OUT R = 1k IN DUT 33 OUT 50 GND1 GND2 RL = 50 1F 33 1F GND2
RL = 50
ZO = 50
Test Circuit 1 Figure 2. Test Circuits 1 and 2
Test Circuit 2
SD00337
NETWORK ANALYZER
5V + V 10F 10F PORT 1
S-PARAMETER TEST SET PORT 2
10F
0.1F
CURRENT PROBE 1mV/mA 16 CAL VCC 33 OUT NC IN DUT 33 OUT 1F GND1 GND2 100 BAL. TRANSFORMER NH0300HB 1F 50 UNBAL. TEST
Test Circuit 3 Figure 3. Test Circuit 3
SD00338
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TEST CIRCUITS (Continued)
NETWORK ANALYZER -5.2V + V
10F
0.1F PORT 1
S-PARAMETER TEST SET PORT 2
10F
0.1F 16
CURRENT PROBE 1mV/mA
CAL GND2 33 OUT
GND1
1F 50 TEST 100 BAL. 1F TRANSFORMER NH0300HB
NC
IN 33 OUT VCC
UNBAL.
Test Circuit 4 Figure 4. Test Circuit 4
SD00339
PULSE GEN.
5V
33 0.1F 1k IN DUT OUT 50 GND1 GND2 OUT 33
1F A ZO = 50 OSCILLOSCOPE B 1F ZO = 50
Measurement done using differential wave forms
Test Circuit 5 Figure 5. Test Circuit 5
SD00545
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TEST CIRCUITS (Continued)
Typical Differential Output Voltage vs Current Input 5V
OUT + IN IIN (A) GND1 GND2 DUT OUT -
+ VOUT (V) -
2.00 1.60 DIFFERENTIAL OUTPUT VOLTAGE (V) 1.20 0.80 0.40 0.00 -0.40 -0.80 -1.20 -1.60 -2.00 -200 -160 -120 -80 -40 0 40 80 120 160 200
CURRENT INPUT (A)
NE5212A TEST CONDITIONS Procedure 1 RT measured at 30A RT = (VO1 - VO2)/(+30A - (-30A)) Where: VO1 Measured at IIN = +30A VO2 Measured at IIN = -30A Procedure 2 Linearity = 1 - ABS((VOA - VOB) / (VO3 - VO4)) Where: VO3 Measured at IIN = +60A VO4 Measured at IIN = -60A V + R T @ () 60mA) ) V OA OB V + R T @ (* 60mA) ) V OB OB
Procedure 3
VOMAX = VO7 - VO8 Where: VO7 Measured at IIN = +130A VO8 Measured at IIN = -130A
Procedure 4
IIN Test Pass Conditions: VO7 - VO5 > 20mV and V06 - VO5 > 20mV Where: VO5 Measured at IIN = +800A VO6 Measured at IIN = -80A VO7 Measured at IIN = +130A VO8 Measured at IIN = -130A
Test Circuit 8 Figure 6. Test Circuit 8
SD00340
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TYPICAL PERFORMANCE CHARACTERISTICS
NE5212A Supply Current vs Temperature
30 INPUT BIAS VOLTAGE (mV) VCC = 5.0V SUPPLY CURRENT (mA) 29 950 900 850 800 750 700 650 25 -60 -40 -20 600 -60 -40 -20 3.25 -60 -40 -20 0 20 40 60 80 100 120 140 OUTPUT BIAS VOLTAGE (V) VCC = 5.0V
NE5212A Input Bias Voltage vs Temperature
NE5212A Output Bias Voltage vs Temperature
3.50 VCC = 5.0V 3.45 PIN 5 3.40 PIN 7 3.35
28
27
26
3.30
0
20 40 60 80 100 120 140
0
20 40 60 80 100 120 140
AMBIENT TEMPERATURE (C)
AMBIENT TEMPERATURE (C)
AMBIENT TEMPERATURE (C)
NE5212A Differential Output Swing vs Temperature
DIFFERENTIAL OUTPUT SWING (V) 3.8 3.6 3.4 3.2 3.0 2.8 2.6 2.4 -60 -40 -20 VCC = 5.0V DC TESTED RL = OUTPUT OFFSET VOLTAGE (mV)
NE5212A Output Offset Voltage vs Temperature
80 60 40 20 0 -20 -40 -60 -60 -40 -20 VCC = 5.0V VOS = VOUT5 - VOUT7
NE5212A Differential Transresistance vs Temperature
DIFFERENTIAL TRANSRESISTANCE (k ) 17.0 16.5 16.0 15.5 15.0 14.5 14.0 -60 -40 -20 0 20 40 60 80 100 120 140 AMBIENT TEMPERATURE (C) VCC = 5.0V DC TESTED RL =
0
20 40 60 80 100 120 140
0
20 40 60 80 100 120 140
AMBIENT TEMPERATURE (C)
AMBIENT TEMPERATURE (C)
NE5212A Power Supply Rejection Ratio vs Temperature
40 POWER SUPPLY REJECTION RATIO (dB) 39 38 37 36 35 34 33 -60 -40 -20 0 20 40 60 80 100 120 140 AMBIENT TEMPERATURE (C) OUTPUT RESISTANCE ( ) VCC = 5.0V VCC = 0.1V DC TESTED OUTPUT REFERRED 17 16 15 14 13 12
NE5212A Output Resistance vs Temperature
NE5212A Typical Bandwidth Distribution (75 Parts from 3 Wafer Lots)
50 PIN 5 SINGLE-ENDED 40 RL = 50 POPULATION (%) 30 20 10 0 112.5 N, F PKG VCC = 5.0V TA = 25C
VCC = 5.0V DC TESTED
PIN 7
PIN 5 11 10 9 -60 -40 -20
0
20 40 60 80 100 120 140
122.5 132.5 142.5 152.5 FREQUENCY (MHz)
162.5
AMBIENT TEMPERATURE (C)
SD00341
Figure 7. Typical Performance Characteristics
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
Gain vs Frequency
12 11 10 9 8 7 6 5 4 3 0.1
Gain vs Frequency
Output Resistance vs Frequency
5.5V OUTPUT RESISTANCE ( )
5.5V 5.0V 4.5V
N PKG PIN 5 TA = 25C
11 10 9 8 7 6 5 4 3 0.1
5.0V 4.5V N PKG PIN 7 TA = 25C
80 70 60 50 40 30 20 10 0.1
GAIN (dB)
GAIN (dB)
N PKG VCC = 5V TA = 25C PIN 5
PIN 7 1 10 FREQUENCY (MHz) 100
1 10 FREQUENCY (MHz)
100
1 10 FREQUENCY (MHz)
100
Output Resistance vs Frequency
100 90 80 70 60 50 40 30 20 10 0.1 OUTPUT RESISTANCE ( ) 11 10 9 GAIN (dB) 8 7 6 5 4 3 1 10 FREQUENCY (MHz) 100 0.1
Gain vs Frequency
Gain vs Frequency
D PKG TA = 25C VCC = 5V
125C -55C
-55C
11 10 9 GAIN (dB) 8 7 6 5 4 3 0.1
125C 25C
85C -55C
-55C N PKG PIN 5 VCC = 5V 1 10 FREQUENCY (MHz)
N PKG PIN 7 VCC = 5V 1 10 FREQUENCY (MHz)
125C
125C 100
100
Gain and Phase Shift vs Frequency
-45 PHASE (o) N PKG PIN 5 VCC = 5V TA = 25C
Gain and Phase Shift vs Frequency
11 10 9 8 7 6 5 4 3 0.1 11 10 9 8 7 6 5 4 3 0.1
Gain and Phase Shift vs Frequency
11 10 9 GAIN (dB) 8 7 6 5 4 3 0.1
-180 GAIN (dB) PHASE (o) D PKG PIN 7 VCC = 5V TA = 25C -270 -360
-180 N PKG PIN 7 VCC = 5V TA = 25C -270 -360 PHASE (o)
-135
-225
1 10 FREQUENCY (MHz)
100
GAIN (dB)
1 10 FREQUENCY (MHz)
100
1 10 FREQUENCY (MHz)
100
SD00342
Figure 8. Typical Performance Characteristics (cont.)
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
Gain and Phase Shift vs Frequency
DIFFERENTIAL OUTPUT VOLTAGE (V) 4.5 11 10 9 GAIN (dB) 8 7 6 5 4 3 0.1 1 10 FREQUENCY (MHz) 100 D PKG PIN 5 VCC = 5V TA = 25C -90 0 PHASE (o )
Output Voltage vs Input Current
DIFFERENTIAL OUTPUT VOLTAGE (V) 125C 85C -55C 25C 2.0
Differential Output Voltage vs Input Current
5.5V 5.0V 4.5V
0
-180
125C 85C -55C25C 2.0 -150.0
5.5V 5.0V 4.5V 0 INPUT CURRENT (A) 150.0
0 INPUT CURRENT (A)
150.0
-2.0 -150.0
Differential Output Voltage vs Input Current
10 2.000 25C -55C OUTPUT VOLTAGE (V) 125C DELAY (ns) 85C 8 6 4 2 0
Group Delay vs Frequency
0
-55C 25C 85C 125C -2.000 -150.0 INPUT CURRENT (A)
150.0
0.1
20
40
60 80 100 120 140 FREQUENCY (MHz)
160
Output Step Response
VCC = 5V TA = 25C 20mV/Div
0
2
4
6
8
10 (ns)
12
14
16
18
20
SD00343
Figure 9. Typical Performance Characteristics (cont.)
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
THEORY OF OPERATION
Transimpedance amplifiers have been widely used as the preamplifier in fiber-optic receivers. The SA5212A is a wide bandwidth (typically 140MHz) transimpedance amplifier designed primarily for input currents requiring a large dynamic range, such as those produced by a laser diode. The maximum input current before output stage clipping occurs at typically 240A. The SA5212A is a bipolar transimpedance amplifier which is current driven at the input and generates a differential voltage signal at the outputs. The forward transfer function is therefore a ratio of the differential output voltage to a given input current with the dimensions of ohms. The main feature of this amplifier is a wideband, low-noise input stage which is desensitized to photodiode capacitance variations. When connected to a photodiode of a few picoFarads, the frequency response will not be degraded significantly. Except for the input stage, the entire signal path is differential to provide improved power-supply rejection and ease of interface to ECL type circuitry. A block diagram of the circuit is shown in Figure 10. The input stage (A1) employs shunt-series feedback to stabilize the current gain of the amplifier. The transresistance of the amplifier from the current source to the emitter of Q3 is approximately the value of the feedback resistor, RF=7k. The gain from the second stage (A2) and emitter followers (A3 and A4) is about two. Therefore, the differential transresistance of the entire amplifier, RT is RT + V OUT(diff) + 2R F + 2(7.2K) + 14.4kW I IN
R IN +
V IN RF + + 7.2K + 103W 70 I IN 1 ) A VOL
More exact calculations would yield a higher value of 110. Thus CIN and RIN will form the dominant pole of the entire amplifier; f *3dB + 1 2p R IN C IN
Assuming typical values for RF = 7.2k, RIN = 110, CIN = 10pF f *3dB + 1 + 145MHz 2p (110) 10 @ 10 *12
The operating point of Q1, Figure 2, has been optimized for the lowest current noise without introducing a second dominant pole in the pass-band. All poles associated with subsequent stages have been kept at sufficiently high enough frequencies to yield an overall single pole response. Although wider bandwidths have been achieved by using a cascade input stage configuration, the present solution has the advantage of a very uniform, highly desensitized frequency response because the Miller effect dominates over the external photodiode and stray capacitances. For example, assuming a source capacitance of 1pF, input stage voltage gain of 70, RIN = 60 then the total input capacitance, CIN = (1+7.5) pF which will lead to only a 12% bandwidth reduction.
The single-ended transresistance of the amplifier is typically 7.2k. The simplified schematic in Figure 11 shows how an input current is converted to a differential output voltage. The amplifier has a single input for current which is referenced to Ground 1. An input current from a laser diode, for example, will be converted into a voltage by the feedback resistor RF. The transistor Q1 provides most of the open loop gain of the circuit, AVOL70. The emitter follower Q2 minimizes loading on Q1. The transistor Q4, resistor R7, and VB1 provide level shifting and interface with the Q15 - Q16 differential pair of the second stage which is biased with an internal reference, VB2. The differential outputs are derived from emitter followers Q11 - Q12 which are biased by constant current sources. The collectors of Q11 - Q12 are bonded to an external pin, VCC2, in order to reduce the feedback to the input stage. The output impedance is about 17 single-ended. For ease of performance evaluation, a 33 resistor is used in series with each output to match to a 50 test system.
OUTPUT + A3
INPUT A1 A2
RF
A4 OUTPUT -
SD00327
Figure 10. SA5212A - Block Diagram
NOISE
Most of the currently installed fiber-optic systems use non-coherent transmission and detect incident optical power. Therefore, receiver noise performance becomes very important. The input stage achieves a low input referred noise current (spectral density) of 3.5pA/Hz. The transresistance configuration assures that the external high value bias resistors often required for photodiode biasing will not contribute to the total noise system noise. The equivalent input RMS noise current is strongly determined by the quiescent current of Q1, the feedback resistor RF, and the bandwidth; however, it is not dependent upon the internal Miller-capacitance. The measured wideband noise was 52nA RMS in a 200MHz bandwidth.
BANDWIDTH CALCULATIONS
The input stage, shown in Figure 12, employs shunt-series feedback to stabilize the current gain of the amplifier. A simplified analysis can determine the performance of the amplifier. The equivalent input capacitance, CIN, in parallel with the source, IS, is approximately 7.5pF, assuming that CS=0 where CS is the external source capacitance. Since the input is driven by a current source the input must have a low input resistance. The input resistance, RIN, is the ratio of the incremental input voltage, VIN, to the corresponding input current, IIN and can be calculated as:
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
VCC1 VCC2 R1 Q2 Q1 R2 GND1 PHOTODIODE R5 R4 GND2 R7 VB2 Q3 R3 Q4 + Q15 R14 Q16 R15 + OUT+ R12 R13 Q11 Q12 OUT-
INPUT
SD00328
Figure 11. Transimpedance Amplifier
VCC R1 INPUT IIN IB Q1 R2 VIN IF VEQ3 IC1 Q2 Q3 R3
No. of incident photons/sec= where P=optical incident power P No. of incident photons/sec = hc l where P = optical incident power P No. of generated electrons/sec = h @ hc l where = quantum efficiency no. of generated electron hole paris no. of incident photons P NI + h @ hc @ e Amps (Coulombs sec.) l where e = electron charge = 1.6 x 10-19 Coulombs h@e Responsivity R = hc Amp/watt l + I + P@R Assuming a data rate of 400 Mbaud (Bandwidth, B=200MHz), the noise parameter Z may be calculated as:1 Z+ I EQ Amp 52 @ 10 *9 + + 1625 Amp qB (1.6 @ 10 *19)(200 @ 10 6)
RF R4
SD00329
Figure 12. Shunt-Series Input Stage
DYNAMIC RANGE
The electrical dynamic range can be defined as the ratio of maximum input current to the peak noise current: Electrical dynamic range, DE, in a 200MHz bandwidth assuming IINMAX = 120A and a wideband noise of IEQ=52nARMS for an external source capacitance of CS = 1pF. DE (Max. input current) + (Peak noise current) (120 @ 10 *6)
D E(dB) + 20 log
where Z is the ratio of RMS noise output to the peak response to a single hole-electron pair. Assuming 100% photodetector quantum efficiency, half mark/half space digital transmission, 850nm lightwave and using Gaussian approximation, the minimum required optical power to achieve 10-9 BER is: P avMIN + 12 hc B Z + 12 (2.3 @ 10 *19) l 200 @ 10 6 1625 + 897nW + * 30.5dBm, where h is Planck's Constant, c is the speed of light, is the wavelength. The minimum input current to the SA5212A, at this input power is: I avMIN + qP avMIN l hc
*9 *19 + 897 @ 10 @ 1.6 @ 10 2.3 @ 10 *19 = 624nA
( 2 52nA) (120mA) D E(dB) + 20 log + 64dB (73nA) In order to calculate the optical dynamic range the incident optical power must be considered. For a given wavelength ; Energy of one Photon = hc watt sec (Joule) l Where h=Planck's Constant = 6.6 x 10-34 Joule sec. c = speed of light = 3 x 108 m/sec c / = optical frequency 12
1998 Oct 07
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
Choosing the maximum peak overload current of IavMAX=120A, the maximum mean optical power is:
VIN
R = 560 IN
OUT- NE5212A OUT+
a. Non-inverting 20dB Amplifier
VIN
R = 560 IN
OUT+ NE5212A OUT-
As with any high-frequency device, some precautions must be observed in order to enjoy reliable performance. The first of these is the use of a well-regulated power supply. The supply must be capable of providing varying amounts of current without significantly changing the voltage level. Proper supply bypassing requires that a good quality 0.1F high-frequency capacitor be inserted between VCC1 and VCC2, preferably a chip capacitor, as close to the package pins as possible. Also, the parallel combination of 0.1F capacitors with 10F tantalum capacitors from each supply, VCC1 and VCC2, to the ground plane should provide adequate decoupling. Some applications may require an RF choke in series with the power supply line. Separate analog and digital ground leads must be maintained and printed circuit board ground plane should be employed whenever possible.
BASIC CONFIGURATION
b. Inverting 20dB Amplifier OUT+ NE5212A OUT-
VIN
R = 560 IN
c. Differential 20dB Amplifier
SD00344
Figure 13. Variable Gain Circuit hcI avMAX 2.3 @ 10 *19(120 @ 10 *6) + lq 1.6 @ 10 *19
A trans resistance amplifier is a current-to-voltage converter. The forward transfer function then is defined as voltage out divided by current in, and is stated in ohms. The lower the source resistance, the higher the gain. The SA5212A has a differential transresistance of 14k typically and a single-ended transresistance of 7k typically. The device has two outputs: inverting and non-inverting. The output voltage in the differential output mode is twice that of the output voltage in the single-ended mode. Although the device can be used without coupling capacitors, more care is required to avoid upsetting the internal bias nodes of the device. Figure 13 shows some basic configurations.
P avMAX +
= 172W or -7.6dBm Thus the optical dynamic range, DO is: DO = PavMAX - PavMIN = -30.5 -(-7.6) = 22.8dB. This represents the maximum limit attainable with the SA5212A operating at 200MHz bandwidth, with a half mark/half space digital transmission at 820nm wavelength.
VARIABLE GAIN
Figure 14 shows a variable gain circuit using the SA5212A and the SA5230 low voltage op amp. This op amp is configured in a non-inverting gain of five. The output drives the gate of the SD210 DMOS FET. The series resistance of the FET changes with this output voltage which in turn changes the gain of the SA5212A. This circuit has a distortion of less than 1% and a 25dB range, from -42.2dBm to -15.9dBm at 50MHz, and a 45dB range, from -60dBm to -14.9dBm at 10MHz with 0 to 1V of control voltage at VCC.
APPLICATION INFORMATION
Package parasitics, particularly ground lead inductances and parasitic capacitances, can significantly degrade the frequency response. Since the SA5212A has differential outputs which can feed back signals to the input by parasitic package or board layout capacitances, both peaking and attenuating type frequency response shaping is possible. Constructing the board layout so that Ground 1 and Ground 2 have very low impedance paths has produced the best results. This was accomplished by adding a ground-plane stripe underneath the device connecting Ground 1, Pins 8-11, and Ground 2, Pins 1 and 2 on opposite ends of the SO14 package. This ground-plane stripe also provides isolation between the output return currents flowing to either VCC2 or Ground 2 and the input photodiode currents to flowing to Ground 1. Without this ground-plane stripe and with large lead inductances on the board, the part may be unstable and oscillate near 800MHz. The easiest way to realize that the part is not functioning normally is to measure the DC voltages at the outputs. If they are not close to their quiescent values of 3.3V (for a 5V supply), then the circuit may be oscillating. Input pin layout necessitates that the photodiode be physically very close to the input and Ground 1. Connecting Pins 3 and 5 to Ground 1 will tend to shield the input but it will also tend to increase the capacitance on the input and slightly reduce the bandwidth.
RFIN 0.1F SD210 IN 51 +5V VCC 0-1V 0-5V NE5212A OUT+ RFOUT OUT-
10k
2.4k
SD00345
Figure 14. Variable Gain Circuit
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
16MHZ CRYSTAL OSCILLATOR
Figure 15 shows a 16MHz crystal oscillator operating in the series resonant mode using the SA5212A. The non-inverting input is fed back to the input of the SA5212A in series with a 2pF capacitor. The output is taken from the inverting output.
DIGITAL FIBER OPTIC RECEIVER
Figures 16 and 17 show a fiber optic receiver using off-the-shelf components. The receiver shown in Figure 16 uses the SA5212A, the Philips Semiconductors 10116 ECL line receiver, and Philips/Amperex BPF31 PIN diode. The circuit is a capacitor-coupled receiver and utilizes positive feedback in the last stage to provide the hysteresis. The amount of hysteresis can be tailored to the individual application by changing the values of the feedback resistors to maintain the desired balance between noise immunity and sensitivity. At room temperature, the circuit operates at 50Mbaud with a BER of 10E-10 and over the automotive temperature range at 40Mbaud with a BER of 10E-9. Higher speed experimental diodes have been used to operate this circuit at 220Mbaud with a BER of 10E-10. Figure 17 depicts a TTL receiver using the SA5212A and the SA5214 fast amplifier system along with the Philips/Amperex PIN diode. The system shown is optimized for 50 Mb/s Non Return to Zero (NRZ) data. A link status indication is provided along with a jamming function when the input level is below a user-programmable threshold level.
+5V
OUT+ NE5212A IN OUT-
SD00346
Figure 15. 16MHz Crystal Oscillator
VEE VCC +5.0 4.7
1.0F
VBB1
VBB1
0.01F 1k 2 0.1F 7 5 0.1F 1k 0.01F 9 10 1 16 7 1/3 10116 11 6
510
510
1k 1k
510
100pF 2 1/3 4 10116 3 8 0.01F 1k 510 0.1F 510 1k 510 5 13 100pF 15 1/3 12 10116 14 ECL ECL
1
NE5212A 68 34
BPF31
-15V 0.01F VEE VBB1 2.7H -5.2V 4.7F NOTE: 1. Tie all VBB points together. 0.1F 4.7F
VBB1
VEE
SD00347
Figure 16. ECL Fiber Optic Receiver
1998 Oct 07
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Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
+VCC 47F C1 C2 .01F
GND
L1 10H
R1 100
R2 220
D1 LED 1 C9 2 100pF 3 4 R3 47k 5 6 7 8 9 LED CPKDET THRESH GNDA FLAG NE5214 JAM VCCD VCCA GNDD TTLOUT IN1B IN1A 20 19 C7 100pF 5 6 7 8 OUT+ GND2 OUT- GND2 NE5212A GND1 GND1 VCC IIN 4 3 2 1
C3 10F
C5 1.0F
.01F C4 .01F
CAZP 18 CAZN 17
C8 0.1F
C6
OUT2B 16 IN8B OUT2A IN8A RHYST 15 14 13 12
BPF31 OPTICAL INPUT
L2 10H
C10 10F
C11 .01F
L3 10H C12 10F
C13 .01F
10
RPKDET 11
R4 5.1k VOUT (TTL)
SD00348
Figure 17. A 50Mb/s TTL Digital Fiber Optic Receiver
1998 Oct 07
15
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
IIN
1
8
GND2
VCC
2
7
OUT-
GND1
3
6
GND2
GND1
4
5
OUT+
ECN No.: 99918 1990 Jul 5
SD00489
Figure 18. SA5212A Bonding Diagram
Die Sales Disclaimer
Due to the limitations in testing high frequency and other parameters at the die level, and the fact that die electrical characteristics may shift after packaging, die electrical parameters are not specified and die are not guaranteed to meet electrical characteristics (including temperature range) as noted in this data sheet which is intended only to specify electrical characteristics for a packaged device. All die are 100% functional with various parametrics tested at the wafer level, at room temperature only (25C), and are guaranteed to be 100% functional as a result of electrical testing to the point of wafer sawing only. Although the most modern processes are utilized for wafer sawing and die pick and place into waffle pack
carriers, it is impossible to guarantee 100% functionality through this process. There is no post waffle pack testing performed on individual die. Since Philips Semiconductors has no control of third party procedures in the handling or packaging of die, Philips Semiconductors assumes no liability for device functionality or performance of the die or systems on any die sales. Although Philips Semiconductors typically realizes a yield of 85% after assembling die into their respective packages, with care customers should achieve a similar yield. However, for the reasons stated above, Philips Semiconductors cannot guarantee this or any other yield on any die sales.
1998 Oct 07
16
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
DIP8: plastic dual in-line package; 8 leads (300 mil)
SOT97-1
1998 Oct 07
17
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
SO8: plastic small outline package; 8 leads; body width 3.9mm
SOT96-1
1998 Oct 07
18
1998 Oct 07 19
0580A
Philips Semiconductors
Transimpedance amplifier (140MHz)
853-0580A 006688
8-PIN (300 mils wide) CERAMIC DUAL IN-LINE (F) PACKAGE
0.055 (1.40) 0.030 (0.76)
0.055 (1.40) 0.030 (0.76)
NOTES: 1. Controlling dimension: Inches. Millimeters are shown in parentheses.
0.303 (7.70) 0.245 (6.22)
-E-
2. Dimension and tolerancing per ANSI Y14. 5M-1982. 3. "T", "D", and "E" are reference datums on the body and include allowance for glass overrun and meniscus on the seal line, and lid to base mismatch. 4. These dimensions measured with the leads constrained to be perpendicular to plane T. 5. Pin numbers start with Pin #1 and continue counterclockwise to Pin #8 when viewed from the top.
PIN # 1 0.100 (2.54) BSC -D- 0.408 (10.36) 0.376 (9.55)
0.070 (1.78) 0.050 (1.27) 0.175 (4.45) 0.145 (3.68)
0.320 (8.13) 0.290 (7.37) (NOTE 4)
-T- SEATING PLANE
0.200 (5.08) 0.165 (4.19)
0.165 (4.19) 0.125 (3.18)
0.035 (0.89) 0.020 (0.51) BSC 0.300 (7.62) (NOTE 4) 0.015 (0.38) 0.010 (0.25) 0.395 (10.03) 0.300 (7.62)
0.023 (0.58) 0.015 (0.38)
T
ED
0.010 (0.254)
Product specification
SA5212A
Philips Semiconductors
Product specification
Transimpedance amplifier (140MHz)
SA5212A
Data sheet status
Data sheet status Objective specification Preliminary specification Product specification Product status Development Qualification Definition [1] This data sheet contains the design target or goal specifications for product development. Specification may change in any manner without notice. This data sheet contains preliminary data, and supplementary data will be published at a later date. Philips Semiconductors reserves the right to make chages at any time without notice in order to improve design and supply the best possible product. This data sheet contains final specifications. Philips Semiconductors reserves the right to make changes at any time without notice in order to improve design and supply the best possible product.
Production
[1] Please consult the most recently issued datasheet before initiating or completing a design.
Definitions
Short-form specification -- The data in a short-form specification is extracted from a full data sheet with the same type number and title. For detailed information see the relevant data sheet or data handbook. Limiting values definition -- Limiting values given are in accordance with the Absolute Maximum Rating System (IEC 134). Stress above one or more of the limiting values may cause permanent damage to the device. These are stress ratings only and operation of the device at these or at any other conditions above those given in the Characteristics sections of the specification is not implied. Exposure to limiting values for extended periods may affect device reliability. Application information -- Applications that are described herein for any of these products are for illustrative purposes only. Philips Semiconductors make no representation or warranty that such applications will be suitable for the specified use without further testing or modification.
Disclaimers
Life support -- These products are not designed for use in life support appliances, devices or systems where malfunction of these products can reasonably be expected to result in personal injury. Philips Semiconductors customers using or selling these products for use in such applications do so at their own risk and agree to fully indemnify Philips Semiconductors for any damages resulting from such application. Right to make changes -- Philips Semiconductors reserves the right to make changes, without notice, in the products, including circuits, standard cells, and/or software, described or contained herein in order to improve design and/or performance. Philips Semiconductors assumes no responsibility or liability for the use of any of these products, conveys no license or title under any patent, copyright, or mask work right to these products, and makes no representations or warranties that these products are free from patent, copyright, or mask work right infringement, unless otherwise specified. Philips Semiconductors 811 East Arques Avenue P.O. Box 3409 Sunnyvale, California 94088-3409 Telephone 800-234-7381 (c) Copyright Philips Electronics North America Corporation 1998 All rights reserved. Printed in U.S.A. Date of release: 10-98 Document order number: 9397 750 04625
Philips Semiconductors
1998 Oct 07 20

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